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LT1249 Power Factor Controller
FEATURES
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DESCRIPTIO
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Standard 8-Pin Packages High Power Factor Over Wide Load Range with Line Current Averaging International Operation Without Switches Instantaneous Overvoltage Protection Minimal Line Current Dead Zone Typical 250A Start-Up Supply Current Rejects Line Switching Noise Synchronization Capability Low Quiescent Current: 9mA Fast 1.5A Peak Current Gate Driver
The 8-pin LT (R)1249 provides active power factor correction for universal offline power systems with very few external parts. By using fixed high frequency PWM current averaging without the need for slope compensation, the LT1249 achieves far lower line current distortion, with a smaller magnetic element than systems that use either peak current detection or zero current switching approach, in both continuous and discontinuous modes of operation. The LT1249 uses a multiplier containing a square gain function from the voltage amplifier to reduce the AC gain at light output load and thus maintains low line current distortion and high system stability. The LT1249 also provides filtering capability to reject line switching noise which can cause instability when fed into the multiplier. Line current dead zone is minimized with low bias voltage at the current input to the multiplier. The LT1249 provides many protection features including peak current limiting and overvoltage protection. The switching frequency is internally set at 100kHz. While the LT1249 simplifies PFC design with minimal parts count, the LT1248 provides flexibilities in switching frequency, overvoltage and current limit.
APPLICATIO S
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Universal Power Factor Corrected Power Supplies Preregulators up to 1500W
, LTC and LT are registered trademarks of Linear Technology Corporation.
BLOCK DIAGRA
VAOUT 5 VSENSE 7.5V 6 IAC 4
+
MOUT 3 RMOUT 4k IA IB 32k MULTIPLIER I 2I IM = A B 2 200A 15A 1V
+
CAOUT 2
EA
-
VCC 16V/10V
+ - -
IM
250A MAX
+
CA
- +
+
gm = 1/3k
RUN 0.7V
M1
- +
SYNC
OSC 20A 35pF
1249 BD
44A 22A
4k
-
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GND 1 7.5V VREF RUN VCC 7 R S Q GTDR 8 16V
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LT1249
ABSOLUTE MAXIMUM RATINGS
Supply Voltage ....................................................... 27V GTDR Current Continuous ..................................... 0.5A GTDR Output Energy (Per Cycle) ............................. 5J IAC Input Current ................................................. 20mA VSENSE Input Voltage ............................................ VMAX MOUT Input Current.............................................. 5mA Operating Junction Temperature Range LT1249C ................................................ 0C to 100C LT1249I ........................................... - 40C to 125C Thermal Resistance (Junction-to-Ambient) N8 Package ................................................ 100C/W S8 Package ................................................. 120C/W Storage Temperature Range ..................-65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
PACKAGE/ORDER INFORMATION
TOP VIEW GND 1 CAOUT 2 MOUT 3 IAC 4 N8 PACKAGE 8-LEAD PDIP S8 PACKAGE 8-LEAD PLASTIC SO
TJMAX = 125C, JA = 100C/W (N8) TJMAX = 125C, JA = 120C/W (S8)
ORDER PART NUMBER
8 7 6 5 GTDR VCC VSENSE VAOUT
LT1249CN8 LT1249IN8 LT1249CS8 LT1249IS8 S8 PART MARKING 1249 1249I
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
PARAMETER Overall Supply Current (VCC in Undervoltage Lockout) Supply Current, On VCC Turn-On Threshold VCC Turn-Off Threshold Voltage Amplifier VSENSE Bias Current Voltage Amp Gain Voltage Amp Unity-Gain Bandwidth Voltage Amp Output High Voltage Amp Output Low Voltage Amp Source Current Voltage Amp Sink Current Threshold Voltage Amp Sink Current Hysteresis Current Amplifier Current Amp Offset Voltage Current Amp Transconductance Current Amp Voltage Gain Current Amp Source Current Current Amp Sink Current Current Amp Output High Current Amp Output Low
The q denotes specifications which apply over the operating temperature range, otherwise specifications are at TA = 25C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, IAC = 100A, CAOUT = 3.5V, VAOUT = 5V, no load on any outputs, unless otherwise noted.
CONDITIONS VCC = Lockout Voltage - 0.2V 11.5V VCC VMAX, CAOUT = 1V
q q q q
MIN
TYP 0.25 8 16.5 10.5 -25 100 1.5 12 0.1 260 44 22.5 2 320 1000 145 95 8.1 1.2
MAX 0.45 12 17.5 11.5 -250
UNITS mA mA V V nA dB MHz V V A A A mV mho V/V A A V V
15.5 9.5
VSENSE = 0V to 7V
q
70 0 Source Current 50A 0 Sink Current 5A Linear Operation, 2V < VAOUT < 10V 2V < VAOUT < 10V
q q q q q q
10 130 33 14
0.4 450 57 30 15 550 220 125 2
ICAOUT = 40A 2.5V VCAOUT 7.5V VMOUT = 1V, IM = 0V VMOUT = - 0.3V, IM = 0A
q
150 500 100 67 7.4
2
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LT1249
ELECTRICAL CHARACTERISTICS
PARAMETER Reference Reference Output Voltage Reference Output Voltage Worst Case Reference Output Voltage Line Regulation Multiplier Multiplier Output Current Multiplier Output Current Offset Multiplier Max Output Current (IM(MAX)) Multiplier Max Output Voltage (IM(MAX) * RMOUT) Multiplier Gain Constant (Note 3) IAC Input Resistance Oscillator Oscillator Frequency Control Pin (CAOUT) Threshold Synchronization Frequency Range Gate Driver Max GTDR Output Voltage GTDR Output High GTDR Output Low (Device Unpowered) GTDR Output Low (Device Active) Peak GTDR Current GTDR Rise and Fall Time GTDR Max Duty Cycle
The q denotes specifications which apply over the operating temperature range, otherwise specifications are at TA = 25C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, IAC = 100A, CAOUT = 3.5V, VAOUT = 5V, no load on any outputs, unless otherwise noted.
CONDITIONS TA = 25C, Measured at VSENSE Pin All Line, Temperature VLOCKOUT < VCC < VMAX IAC = 100A, VAOUT = 5V RAC = 1M from IAC to GND IAC = 450A, VAOUT = 7V (Note 2) IAC = 450A, VAOUT = 7V (Note 2) IAC from 50A to 1mA
q
MIN 7.39 7.32 - 20
TYP 7.5 7.5 5 35 - 0.05 - 250 - 1.1 0.035 32 100 1.8
MAX 7.6 7.68 20
UNITS V V mV A A A V -2 V k kHz V kHz V V V V A ns %
q q
q q q
- 375 - 1.25 15 75 1.3 127 12 VCC - 3.0
- 0.5 - 150 - 0.96 50 125 2.3 160 17.5 1.5 1
Duty Cycle = 0 Synchronizing Pulse Low 0.35V on CAOUT 0mA Load, 18V < VCC < VMAX (Note 4) - 200mA Load, 11.5V VCC 15V VCC = 0V, 50mA Load (Sinking) 200mA Load (Sinking) 10nF from GTDR to GND 1nF from GTDR to GND
q q q q q q
15 0.9 0.5 2 25 96 IM
90 Note 3: Multiplier Gain Constant: K =
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Current amplifier is in linear mode with 0V input common mode.
IAC (VAOUT - 1.5)2
Note 4: Maximum GTDR output voltage is internally clamped for higher VCC voltages.
TYPICAL PERFORMANCE CHARACTERISTICS
Voltage Amplifier Open-Loop Gain and Phase
100 80 GAIN 60
GAIN (dB)
TRANSCONDUCTANCE (mho)
40 20 0 -20
PHASE
10
100
1k 10k 100k FREQUENCY (Hz)
UW
Transconductance of Current Amplifier
0 -20 -40 -60 -80 -100 -120 10M
1249 G01
1249 G02
400 350 300 250 200 150 100 50 0 1k 10k 1M 100k FREQUENCY (Hz) gm
20 0 -20
PHASE (DEG)
-40 -60 -80
PHASE (DEG)
-100 -120 -140 10M
1M
3
LT1249 TYPICAL PERFORMANCE CHARACTERISTICS
Reference Voltage vs Temperature
7.536 7.524
REFERENCE VOLTAGE (V) 300 VAOUT = 6.5V VAOUT = 6V VAOUT = 5.5V VAOUT = 4.5V VAOUT = 4V VAOUT = 3.5V VAOUT = 3V 0 VAOUT = 2.5V VAOUT = 2V 500
1249 G04
7.512 7.500 7.488 7.476 7.464 7.452 7.440 7.428 -75 -50 -25 0 25 50 75 100 125 150 JUNCTION TEMPERATURE (C)
1249 G03
IM (A)
Supply Current vs Supply Voltage
10 9 8 TJ = -55C TJ = 25C
SUPPLY CURRENT (mA)
GTDR VOLTAGE (V)
TJ = 125C
6 5 4 3 2 1 0 10 12 14 16 18 20 22 24 26 28 30 SUPPLY VOLTAGE (V)
1249 G05
16.5 16.0 15.5 15.0 14.5 14.0 13.5 13.0 0 -120 -180 -240 - 60 SOURCE CURRENT (mA) -300
1249 G06
GTDR VOLTAGE (V)
7
GTDR Rise and Fall Time
400 550 500 450
SUPPLY CURRENT (A)
300
TIME (ns)
350 300 250 200 150 100 50 0 -55C 25C 125C
FREQUENCY (kHz)
FALL TIME 200 RISE TIME 100 NOTE: GTDR SLEWS BETWEEN 1V AND 16V 0 0 10 20 30 40 LOAD CAPACITANCE (nF) 50
1249 G08
4
UW
Multiplier Current
VAOUT = 5V
150
0
250 IAC (A)
GTDR Source Current
18.5 18.0 17.5 17.0 VCC = 18V
1.1 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0
GTDR Sink Current
TJ = 125C TJ = 25C TJ = -55C
TA = -55C
TA = 25C TA = 125C 0 60 120 180 240 SINK CURRENT (mA) 300
1249 G07
Start-Up Supply Current vs Supply Voltage
140 130 120 110 100 90 80
Switching Frequency
400
0
2
4
6 8 10 12 14 16 18 20 SUPPLY VOLTAGE (V)
1249 G09
70 25 50 75 -75 -50 -25 0 TEMPERATURE (C)
100 125
1249 G10
LT1249 TYPICAL PERFORMANCE CHARACTERISTICS
Synchronization Threshold at CAOUT
1.0 1.2 1.0 0.8 0.6 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 -25 0 25 50 75 TEMPERATURE (C) 100 125 -1.0 -2.4 0 -1.2 1.2 MOUT VOLTAGE (V) 2.4
1249 G12
SYNCHRONIZATION THRESHOLD (V)
0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 -50
TRANSCONDUCTANCE (mho)
MOUT CURRENT (mA)
Voltage Amp Sink Current Limits (Threshold)
60 50 UP THRESHOLD IM(MAX) x RMOUT (V) 40 30 DOWN THRESHOLD 20 10 0 -75 -50
-1.15 -1.10 -1.05 -1.00 -0.95
DUTY CYCLE (%)
CURRENT (A)
-25 0 25 50 75 TEMPERATURE (C)
NOTE: THESE SINK CURRENT THRESHOLDS ARE FOR OVERVOLTAGE PROTECTION FUNCTION.
PIN FUNCTIONS
GND (Pin 1): Ground. CAOUT (Pin 2): This is the output of the current amplifier that senses and forces the line current to follow the reference signal that comes from the multiplier by commanding the pulse width modulator. When CAOUT is low, the modulator has zero duty cycle. MOUT (Pin 3): The multiplier current goes out of this pin through the 4k resistor RMOUT. The voltage developed across RMOUT is the reference voltage of the current loop and it is limited to 1.1V. The noninverting input of the current amplifier is also tied to RMOUT. In operation, MOUT is normally at negative potential and only AC signals appear at the noninverting input of the current amplifier. IAC (Pin 4): This is the AC line voltage sensing input to the multiplier. It is a current input that is biased at 2V to minimize the crossover dead zone caused by low line voltage. A 32k resistor is in series with the current input, so that a small external capacitor can be used to filter out the switching noise from the high impedance lines. VAOUT (Pin 5): This is the output of the voltage error amplifier. The output is clamped at 12V. When the output goes below 1.5V, the multiplier output current is zero.
UW
1249 G11
MOUT Pin Characteristics
400 125C 25C -50C 350 300 250 200 150 100 50
Transconductance of Current Amplifier Over Temperature
0 -50 -25
0
25
50
75
100
125
TEMPERATURE (C)
1249 G13
Maximum Multiplier Output Voltage (IM(MAX) * RMOUT)
-1.30 -1.25 -1.20
Maximum Duty Cycle
100 99 98 97 96 95 94 93 92 91
100 125
-0.90 -75 -50
-25
0
25
50
75
100 125
1249 G15
90 -50
-25
TEMPERATURE (C)
1249 G14
0 25 50 75 TEMPERATURE (C)
100
125
1249 G16
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LT1249
PIN FUNCTIONS
VSENSE (Pin 6): This is the inverting input to the voltage amplifier. VCC (Pin 7): This is the supply of the chip. The LT1249 has a very fast gate driver required to fast charge high power MOSFET gate capacitance. High current spikes occur during charging. For good supply bypass, a 0.1F ceramic capacitor in parallel with a low ESR electrolytic capacitor, 56F or higher is required in close proximity to IC GND. GTDR (Pin 8): The MOSFET gate driver is a 1.5A fast totem pole output. It is clamped at 15V. Capacitive loads like MOSFET gates may cause overshoot. A gate series resistor of at least 5 will prevent the overshoot.
APPLICATIONS INFORMATION
Error Amplifier The error amplifier has a 100dB DC gain and 1.5MHz unitygain frequency. It is internally clamped at 12V. The noninverting input is tied to the 7.5V reference. Current Amplifier The multiplier output current IM flows out of the MOUT pin through the 4k resistor RMOUT and develops the reference signal to the current loop that is controlled by the current amplifier. Current gain is the ratio of RMOUT to line current sense resistor. The current amplifier is a transconductance amplifier. Typical gm is 320mho and gain is 60dB with no load. The inverting input is internally tied to GND. The noninverting input is tied to the multiplier output. The output is internally clamped at 8V. Output resistance is about 4M; DC loading should be avoided because it will lower the gain and introduce offset voltage at the inputs which becomes a false reference signal to the current loop and can distort line current. Note that in the current averaging operation, high gain at twice the line frequency is necessary to minimize line current distortion. Because CAOUT may need to swing 5V over one line cycle at high line condition, 11mV will be present at the inputs of the current amplifier if gain is rolled off to 450 at 120Hz (1nF in series with 10k at CAOUT). At light load, when (IM)(RMOUT) can be less than 100mV, lower gain will distort the current loop reference signal and line current. If signal gain at the 100kHz switching frequency is too high, the system behaves more like a current mode system and can cause subharmonic oscillation. Therefore, the current amplifier should be compensated to have a gain of less than 15 at 100kHz and more than 300 at 120Hz. Multiplier The multiplier is a current multiplier with high noise immunity in a high power switching environment. The current gain is: IM = (IAC)(IEA2)/(200A)2, and IEA = (VAOUT - 1.5V)/25k With a square function, because of the lower gain at light power load, system stability is maintained and line current distortion caused by the AC ripple fed back to the error amplifier is minimized. Note that switching ripple on the high impedance lines could get into the multiplier from the IAC pin and cause instability. The LT1249 provides an internal 25k resistor in series with the low impedance multiplier current input so that only a capacitor from the IAC pin to GND is needed to filter out the noise. Maximum multiplier output current is limited to 250A. Figure 1 shows the multiplier transfer curves.
300 VAOUT = 6.5V VAOUT = 6V VAOUT = 5.5V VAOUT = 4.5V VAOUT = 4V VAOUT = 3.5V VAOUT = 3V 0 VAOUT = 2.5V VAOUT = 2V 500
1249 G04
IM (A)
6
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VAOUT = 5V
150
0
250 IAC (A)
Figure 1. Multiplier Current IM vs IAC and VAOUT
LT1249
APPLICATIONS INFORMATION
Line Current Limiting Maximum voltage across RMOUT is internally limited to 1.1V. Therefore, line current limit is 1.1V divided by the sense resistor RS. With a 0.2 sense resistor RS line current limit is 5.5A. As a general rule, RS is chosen according RS = (IM(MAX) )(RMOUT )(VLINE(MIN)) K (1.414)POUT(MAX) With ts = 30ns, fs = 130kHz, VC = 3V and R2 = 10k, offset voltage shift is 5mV. Note that this offset voltage will add slight distortion to line current at light load.
CAOUT 1N5712 VCC R1 10k 80pF R2 10k 1nF 2N2369 2k
1249 F02
where POUT(MAX) is the maximum power output and K is usually between 1.1 and 1.3 depending on efficiency and resistor tolerance. When the output is overloaded and line current reaches limit, output voltage VOUT will drop to keep line current constant. System stability is still maintained by the current loop which is controlled by the current amplifier. Further load current increase results in further VOUT drop and clipping of the line current, which degrades power factor. Synchronization The LT1249 can be externally synchronized in a frequency range of 127kHz to 160kHz. Figure 2 shows the synchronizing circuit. Synchronizing occurs when CAOUT pin is pulled below 0.5V with an external transistor and a Schottky diode. The Schottky diode and the 10k pull-up resistor are necessary for the required fast slewing back up to the normal operating voltage on CAOUT after the transistor is turned off. Positive slewing on CAOUT should be faster than the oscillator ramp rate of 0.5V/s. The width of the synchronizing pulse should be under 60ns. The synchronizing pulses introduce an offset voltage on the current amplifier inputs, according to: V - 0.5 (ts)(fs)IC + C R2 VOS = gm ts = pulse width fs = pulse frequency IC = CAOUT source current ( 150A) VC = CAOUT operating voltage (1.8V to 6.8V) R2 = resistor for the midfrequency "zero" in the current loop gm = current amplifier transconductance ( 320mho)
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5V 0V
Figure 2. Synchronizing the LT1249
Overvoltage Protection In Figure 3, R1 and R2 set the regulator output DC level: VOUT = VREF[(R1 + R2)/R2]. With R1 = 1M and R2 = 20k, VOUT is 382V. Because of the slow loop response necessary for power factor correction, output overshoot can occur with sudden load removal or reduction. To protect the power components and output load, the LT1249 voltage error amplifier senses the output voltage and quickly shuts off the current switch when overvoltage occurs. When overshoot occurs on VOUT, the overcurrent from R1 will go through VAOUT because amplifier feedback keeps VSENSE locked at 7.5V. When this overcurrent reaches 44A amplifier sinking limit, the amplifier loses feedback and its output snaps low to turn the multiplier off. Overvoltage trip level: VOUT = (44A)(R1)
0.047F VOUT C1 0.47F
R3 330k
R1 1M
VSENSE
-
VAOUT EA 44A MULTIPLIER LT1249
1249 F03
R2 20k VREF 7.5V
+
22A
Figure 3. Overvoltage Protection
7
LT1249
APPLICATIONS INFORMATION
The Figure 3 circuit therefore has 382V on VOUT, and an overvoltage level = (VOUT + 44V), or 426V. With a 22A hysteresis, VOUT then has to drop 22V to 404V before feedback recovers and the switch turns back on. MOUT is a high impedance current output. In the current loop, offset line current is determined by multiplier offset current and input offset voltage of the current amplifier. A negative 4mV current amplifier VOS translates into 20mA line current and 5W input power for 250V line if 0.2 sense resistor is used. Under no load or when the load power is less than this offset input power, VOUT would slowly charge up to an overvoltage state because the overvoltage comparator can only reduce multiplier output current to zero. This does not guarantee zero output current if the current amplifier has offset. To regulate VOUT under this condition, the amplifier M1 (see Block Diagram), becomes active in the current loop when VAOUT goes down to 1V. The M1 can put out up to 15A to the 4k resistor at the inverting input to cancel the current amplifier negative VOS and keep VOUT error to within 2V. Undervoltage Lockout The LT1249 turns on when VCC is higher than 16V and remains on until VCC falls below 10V, whereupon the chip enters the lockout state. In the lockout state, the LT1249 only draws 250A, the oscillator is off, the VREF and the GTDR pins remain low to keep the power MOSFET off. Start-Up and Supply Voltage The LT1249 draws only 250A before the chip starts at 16V on VCC. To trickle start, a 90k resistor from the power line to VCC supplies the trickle current and C4 holds the VCC up while switching starts (see Figure 4). Then the auxiliary winding takes over and supplies the operating current. Note that D3 and the large value C3, in both Figures 4 and 5, are only necessary for systems that have sudden large load variation down to minimum load and/or very light load conditions. Under these conditions, the loop may exhibit a start/restart mode because switching remains off long enough for C4 to discharge below 10V. The C3 will hold VCC up until switching resumes. For less severe load variations, D3 is replaced with a short and C3 is omitted. The turns ratio between the primary winding and the
LINE MAIN INDUCTOR NP NS D1 R1 90k 1W D3
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D2
VCC C1 2F C2 2F
+
+
C3 390F
+
C4 56F
ALL CAPACITORS ARE RATED 35V
1249 F04
Figure 4. Power Supply for LT1249
C2 1000pF 450V
LINE
MAIN INDUCTOR
D2
D3
R1 90k 1W
+
D1
C3 390F 35V
18V
+
VCC
C4 56F 35V
1249 F05
Figure 5. Power Supply for LT1249
auxiliary winding determines VCC according to: VOUT/(VCC - 2V) = NP/NS. For 382V VOUT and 18V VCC, NP/NS 19. In Figure 5 a new technique for supply voltage eliminates the need for an extra inductor winding. It uses capacitor charge transfer to generate a constant current source which feeds a Zener diode. Current to the Zener is equal to (VOUT - VZ)(C)(f), where VZ is Zener voltage and f is switching frequency. For VOUT = 382V, VZ = 18V, C = 1000pF and f = 100kHz, Zener current will be 36mA. This is enough to operate the LT1249, including the FET gate drive. Output Capacitor The peak-to-peak 120Hz output ripple is determined by: VP-P = (2)(ILOADDC)(Z) where ILOAD DC: DC load current Z: capacitor impedance at 120Hz For 180F at 300W load, ILOADDC = 300W/385V = 0.78A,
LT1249
APPLICATIONS INFORMATION
VP-P = (2)(0.78A)(7.4) = 11.5V. If less ripple is desired, higher capacitance should be used. The selection of the output capacitor should also be based on the operating ripple current through the capacitor. The ripple current can be divided into three major components. The first is at 120Hz whose RMS value is related to the DC load current as follows: I1RMS (0.71)(ILOADDC) The second component contains the PF switching frequency ripple current and its harmonics. Analysis of this ripple is complicated because it is modulated with a 120Hz signal. However, computer numerical integration and Fourier analysis approximate the RMS value reasonably close to the bench measurements. The RMS value is about 0.82A at a typical condition of 120VAC, 200W load. This ripple is line voltage dependent, and the worst case is at low line. I2RMS = 0.82A at 120VAC, 200W The third component is the switching ripple from the load, if the load is a switching regulator. I3RMS ILOADDC For United Chemicon KMH 400V capacitor series, ripple current multiplier for currents at 100kHz is 1.43. The equivalent 120Hz ripple current can then be found: The 120Hz ripple current rating at 105C ambient is 0.95A for the 180F KMH 400V capacitor. The expected life of the output capacitor may be calculated from the thermal stress analysis:
L = (LO )(2)
(105C + TK )-( TAMB + TO ) 10
IRMS =
(I )
1RMS
2
I I + 2RMS + 3RMS 1.43 1.43
2
2
For a typical system that runs at an average load of 200W and 385V output: ILOADDC = 0.52A I1RMS (0.71)(0.52A) = 0.37A I2RMS 0.82A at 120VAC I3RMS ILOADDC = 0.52A
IRMS =
(
0.82A 0.52A 0.37A + + = 0.77A 1.43 1.43
)
2
2
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where L = expected life time LO = hours of load life at rated ripple current and rated ambient temperature TK = capacitor internal temperature rise at rated condition. TK = (I2R)/(KA), where I is the rated current, R is capacitor ESR, and KA is a volume constant. TAMB = operating ambient temperature TO = capacitor internal temperature rise at operating condition In our example, LO = 2000 hours and TK = 10C at rated 0.95A. TO can then be calculated from:
I 0.77A TO = RMS (TK ) = (10C ) = 6.6C 0.95A 0.95A
2 2
Assuming the operating ambient temperature is 60C, the approximate life time is: LO (2000)(2) 57,000 Hrs.
(105C +10C)-(60C + 6.6C) 10
For longer life, capacitor with higher ripple current rating or parallel capacitors should be used. Protection Against Abnormal Current Surge Conditions The LT1249 has an upper limit on the allowed voltage across the current sense resistor. The voltage into the MOUT pin connected to this resistor must not exceed - 6V while the chip is running and -12V under any conditions. The LT1249 gate drive will malfunction if the MOUT pin voltage exceeds - 6V while VCC is powered, destroying the power FET. The 12V absolute limit is imposed by ESD clamps on the MOUT pin. Large currents will flow at
9
LT1249
APPLICATIONS INFORMATION
voltages above 8V and the 12V limit is only for surge conditions. In normal operation, the voltage into MOUT does not exceed 1.1V, but under surge conditions, the voltage could temporarily go higher. To date, no field failures due to surges have been reported for normal LT1249 configurations, but if the possibility exists for extremely large current surges, please read the following discussion. Offline switching power supplies can create large current surges because of the high value storage capacitor used. The surge can be the result of closing the line switch near the peak of the AC line voltage, or because of a large transient in the line itself. These surges are well known in the power supply business, and are normally controlled with a negative temperature coefficient thermistor in series with the rectifier bridge. When power is switched on, the thermistor is cold (high resistance) and surges are limited. Current flow in the thermistor causes it to heat and resistance drops to the point where overall efficiency loss in the resistor is acceptable. This basic protection mechanism can be partially defeated if the power supply is switched off for a few seconds, then turned back on. The thermistor has not had time to cool significantly and if the subsequent turn-on catches the AC line near its peak, the resulting surge is much higher than normal. Even if this surge current generates a voltage greater than 6V (but less than 12V) across the sense
THERMISTOR
+
BRIDGE SURGE PATH RS 100 MOUT LT1249
-
Figure 6. Protecting MOUT from Extremely High Current Surges
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resistor, the standard LT1249 application will not be affected because the chip is not yet powered. Problems are only created if the VCC pin is powered from some external housekeeping supply that remains powered when bridge power is switched off. A huge line voltage surge, beyond the normal worst-case limits, can also create a large current surge. The peak of the line voltage must significantly exceed the storage capacitor voltage (typically 380V) for this to occur, so peak line voltage would probably have to exceed 450V. Such excessive surges might occur if a very large mains load was suddenly removed, with a resulting line "kickback". If the surge results in voltage at the MOUT pin greater than 6V, it must also last more than 30s (three switch cycles) to cause FET problems. External Clamp The external clamp shown in Figure 6 will protect the LT1249 MOUT pin against extremely large line current surges (see above). Protection is provided for all VCC power methods. The 100 resistor and three diodes limit the peak negative voltage into MOUT to less than 3V. Current sense gain is attenuated by only 100/4000 = 2.5%. Three diodes are used because the peak negative voltage into MOUT in normal operation could go as high as -1.1V and the diodes should not conduct more than a few microamps under this condition.
+
STORAGE CAPACITOR
LT1249
PACKAGE DESCRIPTION
0.300 - 0.325 (7.620 - 8.255)
0.009 - 0.015 (0.229 - 0.381)
(
+0.035 0.325 -0.015 8.255 +0.889 -0.381
)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0- 8 TYP
0.014 - 0.019 (0.355 - 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.016 - 0.050 (0.406 - 1.270)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400* (10.160) MAX 8 7 6 5
0.255 0.015* (6.477 0.381)
1
2
3
4 0.130 0.005 (3.302 0.127)
0.045 - 0.065 (1.143 - 1.651)
0.065 (1.651) TYP 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 0.003 (0.457 0.076)
N8 1098
0.100 (2.54) BSC
S8 Package 8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 - 0.197* (4.801 - 5.004) 8 7 6 5
0.228 - 0.244 (5.791 - 6.197)
0.150 - 0.157** (3.810 - 3.988)
1
2
3
4
0.053 - 0.069 (1.346 - 1.752)
0.004 - 0.010 (0.101 - 0.254)
0.050 (1.270) BSC
SO8 1298
11
LT1249
TYPICAL APPLICATION
750H* 90V TO 270V MURH860 VOUT
+
6A EMI FILTER
-
IRF840 1M
0.047F
0.47F
330k
5
+
VAOUT
7.5V 6 1M 4 IAC VSENSE
EA
-
IA IB 32k
15A 1V
+
gm = 1/3k
M1
+ -
4.7nF
44A 22A
4k
* 1. COILTRONICS CTX02-12236 (TYPE 52 CORE) AIR MOVEMENT NEEDED AT POWER LEVEL GREATER THAN 250W. 2. COILTRONICS CTX02-12295 (MAGNETICS Kool M(R) 77930 CORE) ** THIS SCHOTTKY DIODE IS TO CLAMP GTDR WHEN MOS SWITCH TURNS OFF. PARASITIC INDUCTANCE AND GATE CAPACITANCE MAY TURN ON CHIP SUBSTRATE DIODE AND CAUSE ERRATIC OPERATIONS IF GTDR IS NOT CLAMPED. SEE APPLICATIONS INFORMATION SECTION FOR CIRCUITRY TO SUPPLY POWER TO VCC.
RELATED PARTS
PART NUMBER LT1103 LT1248 LT1508 LT1509 DESCRIPTION Off-Line Switching Regulator Full Feature Average Current Mode Power Factor Controller Power Factor and PWM Controller Power Factor and PWM Controller COMMENTS Universal Off-Line Inputs with Outputs to 100W Provides All Features in 16-Lead Package Simplified PFC Design Complete Solution for Universal Off-Line Switching Power Supplies
Kool M is a registered trademark of Magnetics, Inc.
12
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
-
-
+
U
+
180F
RS 0.2 1nF 10k MOUT RMOUT 4k MULTIPLIER I 2I IM = A B 2 200A MAX 250A IM CA
+
20k
100pF
10
3
2
CAOUT
1
GND 7 7.5V VREF
V
CC
VCC 16V/10V
+ - - +
RUN
R RUN S
Q 8 GTDR
0.7V
-
SYNC
OSC 20A 35pF 16V ** 1N5819
1249 TA01
1249fb LT/TP 0799 2K REV B * PRINTED IN USA
(c) LINEAR TECHNOLOGY CORPORATION 1994

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